Dual-band dual-polarized sub-6 GHz phased array antenna with suppressed higher order modes

This manuscript proposes a dual-band dual-polarized 64-element phased array antenna (PAA) with suppressed higher order modes (HoMs). Each array element is a microstrip patch structure with complementary split ring resonator (CSRR) loading, allowing for simultaneous dual-band and dual-polarization performance at 3.45 GHz and 3.87 GHz. The proposed PAA exhibits impedance matching of \documentclass[12pt]{minimal} \usepackage{amsmath} \usepackage{wasysym} \usepackage{amsfonts} \usepackage{amssymb} \usepackage{amsbsy} \usepackage{mathrsfs} \usepackage{upgreek} \setlength{\oddsidemargin}{-69pt} \begin{document}$$\le - 20$$\end{document}≤-20 dB at 3.45 GHz and 3.87 GHz while limiting the mutual coupling below \documentclass[12pt]{minimal} \usepackage{amsmath} \usepackage{wasysym} \usepackage{amsfonts} \usepackage{amssymb} \usepackage{amsbsy} \usepackage{mathrsfs} \usepackage{upgreek} \setlength{\oddsidemargin}{-69pt} \begin{document}$$- 19$$\end{document}-19 dB between the adjacent elements at both the operating bands. The proposed design includes a plated through hole placed near the CSRR loading to suppress any higher-order modes (HOMs). Two antenna prototypes, without and with HOMs suppression techniques are fabricated and measured. In addition, we demonstrate the beam steering of the proposed PAA up to \documentclass[12pt]{minimal} \usepackage{amsmath} \usepackage{wasysym} \usepackage{amsfonts} \usepackage{amssymb} \usepackage{amsbsy} \usepackage{mathrsfs} \usepackage{upgreek} \setlength{\oddsidemargin}{-69pt} \begin{document}$$\pm 60^0$$\end{document}±600 using an off-the-shelf 6-bit phase shifter module. The proposed 64 -element array achieves a gain of 23 dBi, and shows a minimal steering loss of \documentclass[12pt]{minimal} \usepackage{amsmath} \usepackage{wasysym} \usepackage{amsfonts} \usepackage{amssymb} \usepackage{amsbsy} \usepackage{mathrsfs} \usepackage{upgreek} \setlength{\oddsidemargin}{-69pt} \begin{document}$$\simeq 3$$\end{document}≃3 dB over the steering angle.

The future wireless communication technology is expected to use intelligent metasurfaces, massive beamforming for improved user experience, potentially providing dedicated links to individual users [1][2][3][4][5] .This communication technology can be fulfilled using a phased array antenna (PAA), which provides multiple beams with highspeed and precise beam forming, beam steering in azimuth, and elevation plane over a wide angular coverage 6,7 .The mmWave beam forming technique using PAA is discussed in [8][9][10] .However, because of strict line-of-sight (LOS), prone to signal degradation due to natural obstructions like trees, buildings, and poor signal penetration capabilities in the mm-wave frequency band (FR2), researchers in academia and industry are focused on the realization of base station antenna in FR1 frequency band 11,12 .Several works discuss the 5G base station antenna using the phased array technique, 13,14 , operating at single resonance and single polarized.However, the upcoming frequency standards in future communication technology, such as 5G require multi-band PAAs, which can save cost and space for additional base stations 15 .On the other hand, the multi-path fading in the base station communication system is a critical parameter to study, which can be addressed using the pattern diversity method with a dual-polarization 16 .
A dual-band dual-polarized antenna using modified mushroom unit cell 17 , double loop dipole antenna with metasurface 18 , differential-fed with a microstrip line with open-circuit stepped-impedance resonators 19 , a massive metamaterial loaded MIMO antenna 20 for sub-6 GHz base station applications are discussed.In sub-6 GHz, a 4 × 4 array configuration using wideband spiral antenna demonstrates beam steering of ±40 021 .Few designs report the baffles shape with FSS and ferrite loading 22 , bent dipole embedded with two individual microstrip balun 23 , produces dual-band, dual-polarized 5G base station arrays.A butler matrix embedded antenna array 24 , demonstrates fixed beam steering up to ±40 0 .In 25 , a dual-band dual-polarized multi-beam antenna configura- tion is demonstrated for ±40 0 fixed beam steering.A dual-polarized 10 × 10 PAA using a cavity-backed patch embedded with vertical orthogonal balun shows beam steering up to ±45 0 is discussed in 26 .However, the cavity- like structure of microstrip antenna is very common for higher order modes (HOMs) excitation.The HOMs accelerate the EMI/EMC issues and give rise to spurious harmonic generation in the RF and communication system of the base station unit leading to radiation pattern distortion, and undesirable side lobe level (SLL).In addition to these unwanted problems, due to energy distribution in the unwanted HOMs, the main beam gain of the array decreases when the main beam of the phased array scans to the larger angles.Moreover, when the unwanted SLL is more, the array can not scan the large angle 27 .To the best of the author's knowledge, a dualband dual-polarized massive PAA with wide-angle beam steering and suppressed HOMs for sub-6 GHz is not available in open literature.
This article proposes a massive ( 8 × 8 ) dual-band, dual-polarized PAA using CSRR loading for sub-6 GHz 5G base station applications, where a plated through hole (PTH) is used to suppress the HOMs while improving the impedance matching.Further, the beam steering performance is demonstrated up to ±60 0 using Ansys HFSS Electromagnetic solver as well as in measurement using a 6−bit phase shifter module.

Methods
Design of dual-band dual-polarized 8 × 8 PAA with reduced HOM The proposed dual-band dual-polarized 8 × 8 antenna is designed on a single-layer dielectric substrate (Rog- ers 3003), with ǫ r = 3 , tanδ = 0.001 and 1.52 mm thickness.Here, an individual antenna element is a square microstrip patch antenna ( L p × L p ) with CSRR loading (see Fig. 1).The CSRR has a dimension of p × p with a split gap (g) and ring width of (r) and is etched out from the top surface of the antenna radiator as shown in Fig. 1b.The CSRR at the top surface is a high Q-factor resonator, which exhibits capacitive and magnetic coupling through the ring slot and split gap respectively.This coupled field between the CSRR and patch exhibits dual-band response without affecting primary radiation pattern 28,29 .Next, the excitation and CSRR positions are optimized thoroughly in the HFSS full-wave solver (see Fig. 2).The S-parameter plot, in Fig. 3a, shows the dual-band response at 3.45 GHz and 3.87 GHz of the proposed design.In order to achieve proper functionality of PAA with low mutual coupling, the antennas are spaced at a distance of 0.5 0 , where 0 is the free-space wavelength at 3.87 GHz operating frequency.This spacing allows for mutual coupling between adjacent elements to be less than or equal to −19 dB, with impedance matching of less than or equal to −20 dB at both resonances (see Fig. 3a).The polarization state of the primary CSRR-loaded square patch element (i.e.m = n = 0 ) in 8 × 8 PAA is described by the simulated surface current distribution at two operating frequencies (see Fig. 3b,c).The proposed antenna has mutually orthogonal field directions at φ = 135 0 and φ = 45 0 for 3.45 GHz and 3.87 GHz, representing a dual polarization state (refer to Fig. 3b,c).In addition to the dual-band and dual-polarization  state, the proposed antenna loaded with CSRRs also exhibits undesired higher-order modes, as shown in the S-parameter response in Fig. 3a.

Configuration of dual-band massive antenna array with HOMs suppression technique
In the proposed microstrip antenna loaded with CSRR, the unwanted higher-order mode (HoM) is suppressed by placing a PTH near the CSRR.The location of the PTH is determined through a parametric study, as demonstrated in the S-parameter plot shown in Fig. 4a.Figures 4b,c display the simulated surface current distribution along φ = 135 0 and φ = 45 0 at 3.45 GHz and 3.87 GHz, respectively, for the CSRR-loaded antenna in the pres- ence of PTH.The presence of PTH does not interrupt operating frequencies or surface current distribution.The PTH suppresses the HOMs present in the X-band region ( 4 − 9 GHz) while improving the dual-band impedance matching (see Fig. 4a).In order to provide a clearer insight into HOMs suppression, a simplified approximate circuit model [28][29][30] of the patch with CSRR loading and CSRR loaded patch with via is presented in Fig. 5a,b respectively.Here, the co-axial excitation is represented as the inductive probe ( L f ).The patch is represented as the RLC resonator ( R 1 , L 1 , C 1 ) and the CSRR loading is represented as a shunt RLC resonator ( R 2 , L 2 , C 2 ) which exhibits a low resonance frequency compared to conventional patch shown in Fig. 1a.Similarly, the HOMs of the patch and CSRR are represented as separate RLC resonators ( ), as shown in Fig. 5a, responsible for the unwanted auxiliary radiation.Figure 5b presents a CSRR-loaded patch with a via ( L 3 ), which routes the fundamental radiating mode current through a short circuit path to ground thereby suppressing the HOMs.
Next, the element with suppressed HOMs is used to configure 8 × 8 antenna array with an inter-element spacing of 0.5 0 .The S-parameter response of the 8 × 8 antenna array is studied in Fig. 6.It is observed that the mutual coupling between antenna elements in the array is below −20 dB, regardless of the antenna position in the array (see Fig. 6).

PAA beam steering performance
The 5G base station technology demonstrates three sector deployment structure with 3D sectorization of the beam pattern in azimuth and elevation, where each sector is equipped with a beam-forming antenna module (usually a PAA) to provide an angular range of 120 0 (i.e ±60 0 ) 31,32 .In this scenario, the PAA steers the beam by controlling the aperture distribution.This is achieved by compensating the progressive phase shift in the array factor (AF) of the planar M × N array and represented in Eq. (1) 33 .
where, I 0 is the uniform excitation amplitude for all m and n elements, k 0 is the wave number.The common main beam of PAA is in the direction of θ = θ 0 is the beam steering angle and φ = φ 0 is the plane of the beam steering.The principal maximum of the main beam is specified by (θ 0 , φ 0 ) .To steer the main beam from the broadside, the element steering phase ( β x , β y ) is summed at each element (m, n) for providing a combined phasing ( ph ) at the mn th element using Eqs.( 2)-(4).Moreover, the steered beam is controlled by active element pattern 33,34 .
The active element pattern can be reshaped by controlling the mutual coupling between the elements.This is possible with increasing the inter-element spacing while maintaining the maximum peak gain and good side lobe levels, but large spacing raises the grating lobe issues.The element spacing is a trade-off between the grating lobe and mutual coupling issues in PAA 35 .For beam steering upto ±60 0 , the PAA must satisfy the Eq. ( 5) at the (1) AF = I 0 � M m=1 e j(m−1)(k 0 d x sin θ cos φ+β x ) × � N n=1 e j(n−1)(k 0 d y sin θ sin φ+β y ) (2)  highest operating frequency to confirm a single main lobe and no grating lobe in −π/2 < θ < π/2 region.Fur- ther, the PAA with N elements restricts the Eq. ( 5) by (N − 1)/N and the 3 dB beam width of the antenna array 34 .
Here, the proposed 8 × 8 PAA uses 0.5 0 spacing to minimize the mutual coupling up to ≤ −19 dB and achieves a beam steering up to ±60 0 .The beam steering performance is demonstrated using HFSS EM solver by exciting each element with a uniform amplitude and phase coefficient (using Eq. 4) [36][37][38] .The solver calculates the absolute phase coefficient for each element and steers the main beam to the desired angle.The 3D beam steering pattern is presented using contour plots at 30 0 step (see Fig. 7).A minimum steering loss close to 3 dB is achieved by optimizing the active element for a stable radiation pattern on the order of cos 1.25 θ 0 39 .In addition to this, the proposed 8 × 8 PAA shows a consistent active return loss of ≤ −12 dB (see Fig. 8), which ensures no scanning blindness during the beam steering performance up to ±60 0 .

Experimental characterization of dual-band dual-polarized 8 × 8 PAA with reduced HOM
The proposed massive array without and with HOMs suppression is fabricated.The fabricated PCB and the co-axial connector (SMA-F) are assembled on a metal plate.First, the fabricated prototype is validated through S-parameter measurement (see the test setup in Fig. 9a,b).Here, the measurement is conducted using Keysight PNA.The port-1 and port-2 of the PNA are connected to port-1 and port-2 of the AUT while terminating the remaining ports with 50 load.The measured S-parameter results show both the prototypes achieve dual-band (5)    resonance with impedance matching of ≤ −20 dB and mutual coupling of ≤ −19 dB and are in good agreement with the simulated S-parameter result (see Fig. 10a).In addition to this, the total active reflection coefficient (TARC) of ≤ −15 dB is achieved at both the operating frequency (see Fig. 10b), which assure efficient operation in active mode 28 .
Next, the radiation pattern of the proposed CSRR-loaded massive antenna array is measured in the near field test range (NFTR).The proposed antenna array is integrated with an RF phase shifter and control module with RF cables (see Fig. 9c).The phase shifter module incorporates a 1 : 32 way power divider (PD) network, where each output port of PD has a 6-bit digital phase shifter and RF switch to enable phase control of each element, which is commercial off-the-shelf hardware of M/s Astra microwave product ltd (2 phase shifter modules are used here).The integrated RF module introduces phase errors, which are captured in discrete measurements.The phase errors are normalized using the phase collimation process, where a suitable phase state is applied to  each element.First, the boresight un-calibrated pattern is measured.The un-calibrated pattern is not capable of producing a perfect SLL and first null, which is essential to estimate the Rayleigh resolution.Later, the antenna array is calibrated using the phase collimation process.In this, the proposed antenna array plane is transformed to measurement plane in the NFTR acquisition system.The NFTR probe is aligned to the antenna elements as per the measurement plane.The respective phase error of each element is acquired and then corrected by compensating the suitable phase state through the control and phase shifter module 40,41 .This process is repeated for both the resonances to ensure nearly zero beam pointing error and minimum SLL.The calibrated antenna array yields a well-defined SLL ( ≃ 13 dB) and first null (see Fig. 11a,b).The proposed antenna achieves boresight gain of 21.8 dBi, 23 dBi, and 3 dB beam width ( θ b ) of 14 0 , 12.5 0 at 3.45 GHz, 3.87 GHz respectively and are in good  agreement with simulated results.Next, the beam steering patterns are measured at every 15 0 step (see Fig. 11c,d), by loading the required phase state to the phase shifter module (using equ.4).It is observed, at θ 0 = 60 0 the 3 dB beam width is increased to 24 0 , 22 0 at 3.45 GHz, 3.87 GHz respectively (with a factor of θ b / cos θ 0 ).Also, the steered beam located at ≈ 56 0 , ≈ 53 0 (i.e ≤ 60 0 ) at 3.45 GHz, 3.87 GHz respectively, which is because of the active element pattern with a factor of √ cos θ and applied progressive phase bits of �β = 5.625 0 corresponds to the 6-bit phase shifter.A comparison study between the available literature and the proposed sub-6 GHz antenna array is presented in Table I.It infers that, a dual-band dual-polarized wide angle beam steering up to ±60 0 ( ≈ 56 0 , ≈ 53 0 measured) with suppressed HoMs.This shows the superiority of the proposed work compared to published works.

Conclusion
Here, a 64 element PAA configuration uses a CSRR-loaded antenna element, which facilitates achieving a wideangle beam steering performance at both resonances.Here, a CSRR loading on a square microstrip antenna is studied for dual-band dual-polarized operation at 3.45 GHz, 3.87 GHz, with improved impedance matching ≤ −20 dB and mutual coupling ≤ −19 dB.The HOMs suppression is achieved by placing a PTH near the vicinity of the CSRR loading.To date, fixed beam steering using Butler matrix circuit-based antennas is demonstrated 24,25 , even a modified cavity technique is reported to enhance the scan performance 26 by reducing the mutual coupling, and it is limited up to ±45 0 by the element size.The proposed array configuration uses a CSRR-loaded antenna element, which facilitates achieving a wide angle beam steering performance up to ≈ 56 0 , ≈ 53 0 at 3.45 GHz, 3.87 GHz respectively.The wide angle beam steering performance is studied using a contour plot and validated by measuring the fabricated prototype.The phase collimation method helps to achieve a well-defined SLL and first null.The proposed antenna array achieves a maximum gain of 23 dBi, while maintaining a minimum scan loss of ≃ 3 dB.

Figure 2 .
Figure 2. Sensitivity analysis of the CSRR Loaded patch antenna (a) by changing the excitation position (b) by changing the CSRR location (c) single element radiation pattern w/o and with CSRR loading.

Figure 5 .
Figure 5. Equivalent circuit model of (a) patch with CSRR loading, (b) patch with CSRR loading and PTH.

Figure 8 .
Figure 8. Active return loss of the 8 × 8 PAA during beam steering analysis.

Figure 10 .
Figure 10.(a) Comparison of S-parameter result of 8 × 8 PAA without and with HOMs suppression, (b) comparison of TARC result.

Table 1 .
Comparison of the proposed antenna array with available open literature.